The disclosed embodiments relate to low noise amplifiers (LNAs).
2. Background Information
The first amplification stage in a radio receiver such as a receiver of a cellular telephone is generally an amplifier circuit called a Low Noise Amplifier (LNA). Measures of LNA operational performance include the noise factor (F) of the LNA and the linearity of the LNA.
The receiver of a cellular telephone includes what is referred to as a receive chain. The receive chain involves a LNA, that outputs a signal to a mixer. The mixer in turn outputs a signal to a baseband filter. The noise factor (F) of the overall receive chain to a first approximation is equal to the noise factor of the LNA plus a quantity, where the quantity is the noise factor of the following stages (the mixer and the baseband filter) divided by the gain of the LNA. Increasing the gain of the LNA therefore decreases the noise factor of the overall receive chain. In a cellular telephone application, there typically are noise factor requirements imposed on the overall receiver. Accordingly, the LNA in a cellular telephone has to have adequate gain to meet noise factor requirements of the overall receiver.
An amplifier, such as the LNA, exhibits an amount of non-linearity. If an ideal sinusoidal input signal of the pure single frequency were supplied to the input of a linear amplifier, then the amplifier would output an amplified version of the input signal. The output signal would have only a single frequency, and this frequency will be the frequency of the input signal. If, however, the same sinusoidal input signal were supplied to the input of an amplifier that exhibits an amount of non-linearity, then the amplifier would output an amplified version of the input signal at the frequency of the input signal, but the amplifier would also output one or more other signals of other frequencies. These other signals are referred to as “distortion”. In a practical receiver, these distortion components are often far away from the frequency of the desired signal and can therefore be filtered out of the receiver output signal. If, however, there is another noise signal (referred to here as a jammer) that is received along with the desired signal into the input of the amplifier, then a complex type of distortion sometimes referred to as cross-modulation distortion can occur. Because this cross-modulation distortion may be close in frequency of the frequency of the desired signal, it is difficult or impossible to filter the cross-modulation distortion out of the receiver output signal. If the cross-modulation distortion components cannot be removed from the output signal by filtering, then the amplifier is made to be more linear so that the magnitude of the cross-modulation distortion components is an acceptable amount.
This requirement to have good linearity may, however, only be imposed when the receiver is operating in the presence of a jammer. If it is known that there is no jammer present, then the linearity requirement on the amplifier can be relaxed without the receiver output signal having an unacceptably large amount of distortion because there will be no cross-modulation generated. For example, in some radio communication protocols, the transmitter may be transmitting at the same time that the receiver is receiving. The frequencies of the transmitted signals are close in frequency to the frequencies of the signals being received. Due to the physical proximity of the transmitter and receiver in the cellular telephone handset, and due to the power of the transmitted signal, some of the transmitted signal may leak back into the receiver and constitute a jammer. This particular jammer is, however, only present when the transmitter is transmitting. When the transmitter is not transmitting, the cross-modulation distortion problem is less severe or absent and the linearity requirements on the receiver can be relaxed. In many LNA topologies, the linearity of the amplifier can be increased by increasing the bias current flowing through the LNA. Similarly, the linearity of the amplifier can be reduced by reducing the bias current flowing through the LNA.
FIG. 1 (Prior Art) is a circuit diagram of one particular differential LNA 1 that utilizes the Post-Distortion Cancellation technique (sometimes referred to as the Active Post-Distortion technique). This technique involves the use of four field effect transistors (FETs) 2-5 biased in the saturation region. FETs 2 and 3 are referred to as the main FETs. FETs 4 and 5 are referred to as the cancel FETs. The left-hand pair of main FET 2 and cancel FET 4 operates as follows. Main FET 2 amplifies an input signal received on input lead 6. An amplified version of the input signal is generated onto node 6. Because main FET 2 is configured as a common source amplifier, the amplified signal has a phase shift of approximately 180 degrees with respect to the input signal on input lead 5. Distortion components are also present in the signal on node 6 along with the desired amplified version of the input signal. The phase-shifted signal on node 6 is applied to the gate input of cancel FET 4. Cancel FET 4 is also biased in the saturation region, but it is designed to be a lousy amplifier in that it generates comparatively more of the distortion components in comparison to amplified desired signal than does main FET 2. Due to the way cancel FET 4 receives its input signal, the phase of the input signal supplied to cancel FET 4 is 180 degrees out of phase with respect to the of the input signal supplied to main FET 2. Accordingly, the desired amplified signal as output from cancel FET 4 is 180 degrees out of phase with respect to the desired amplified signal as output from main FET 2, and the phase of the distortion as output from cancel FET 4 is also 180 degrees out of phase with respect to the distortion as output from main FET 2. The signals output from main FET 2 and cancel FET 4 are summed on a merging node 7. If the magnitude of the distortion output by cancel FET 4 is set to be equal in magnitude to the distortion output by main FET 2, then the distortion signals will cancel each other on node 7. At the same time, some of the desired signal output by main FET 2 will be cancelled by the desired signal output by cancel FET 4, but due to the fact that cancel FET 4 is a lousy amplifier some of the desired signal as output from main FET 2 will remain on node 7. This remaining desired signal is the signal output from the PDC LNA. The other complementary pair of main and cancel FETs 3 and 5 works in a similar fashion. Unfortunately, the cancellation of some of the desired signal on the merging nodes 7 and 8 reduces the gain of the PDC LNA.
The PDC LNA of FIG. 1 has a high linearity mode and a low linearity mode. In the high linearity mode, a bias circuit increases the bias voltage on the gates of main FETs 2 and 3. This increases the DC bias current in the LNA and improves linearity. In the low linearity mode, the bias circuit decreases the bias voltage on the gates of main FETs 2 and 3, thereby degrading linearity somewhat but advantageously reducing power consumption. For further details on the active post-distortion cancellation LNA, see: 1) Published U.S. Patent Application No. 2007/0229154, published Oct. 4, 2007, and 2) Published U.S. Patent Application No. 2007/0030076, published Feb. 8, 2007. The input capacitance of the LNA of FIG. 1 is advantageously low because the gate of only one transistor is coupled to each of the input leads 5 and 9. Unfortunately, PDC LNA 1 has less than optimal gain performance due to the cancel transistors canceling some of the desired signals as output by the main transistors.
FIG. 2 is a circuit diagram of another differential LNA 10 that utilizes a variant of the Derivative Super-position (DS) technique referred by here as the Cross-Coupled Modified Derivative Super-position technique (CCMDS). In this circuit, the main FETs 11-14 are biased in the saturation region, but cancel transistors 15 and 16 are biased in the sub-threshold region. When the transconductance equation that describes the output current of a FET amplifier whose FET is biased in the saturation region is compared to the transconductance equation for a FET amplifier whose FET is biased in the sub-threshold region, it is recognized that the signs of the third order coefficients of the transconductance equations of the two transistors are opposite one another. The signs of the first order coefficients, however, are not opposite one another. In the circuit of FIG. 2, this means that biasing a transistor in the sub-threshold region results in a shift in the phase of the third order distortion it outputs as compared to a transistor biased in the saturation region, whereas the phase of the desired signal as output by the sub-threshold biased transistor is not phase shifted as compared to the transistor biased in the saturation region. The current output by cancel FET 15 is supplied onto merging node 17 such that the phase of the desired signal as output by cancel FET 15 is in phase with the desired signal as output by main FET 11. Because the phase of the third order distortion components as output by cancel FET 15 are 180 degrees out of phase with respect to the amplified desired signal as output by cancel FET 15, the third order distortion components as output by cancel FET 15 are 180 degrees out of phase with respect to the third order distortion components as output by main FET 11. If the magnitudes of the third order distortion components in the cancel and main signal paths are set appropriately, then the third order distortion components on merging node 17 will cancel each other. Advantageously, because the phase of the amplified versions of the desired signal as output by the main and cancel FETs are in phase with respect to one another, both the main FET 11 and the cancel FET 15 work together to amplify the desired signal. The CCMDS LNA of FIG. 2 therefore has improved gain characteristics as compared to the post-distortion LNA of FIG. 1.
The CCMDS LNA of FIG. 2 is operable in two modes. A bias circuit controls the DC bias voltage on the gates of the main FETs 11-14. It controls the gate biases such that either transistors 11 and 12 are operating as the main FETs, or such that transistors 13 and 14 are operating as the main FETs. In a high linearity mode, transistors 11 and 12 are employed as the main FETs and transistors 13 and 14 are disabled. The capacitors 19 and 20 capacitively couple the receiver inputs 21 and 22 to the gates of the main transistors 11 and 12, respectively, and operate as capacitive voltage dividers. The input signal received on the inputs 21 and 22 is therefore attenuated so less of the jammer is supplied onto the gates of main FETs 11 and 12. The main FETs 11 and 12 are biased with a higher bias current so that the strong jammer signal will not cause large signal swings in the amplifier and generate more distortion.
In a low linearity mode, transistors 13 and 14 are employed as the main FETs and transistors 11 and 12 are disabled. Capacitors 19 and 20 are not in the signal path. Because there is no strong jammer present to cause the large signal swings in the amplifier that generate more distortion, main FETs 13 and 14 can be biased at lower bias currents in the low linearity mode than main FETs 11 and 12 are biased in the high linearity mode.
Although the CCMDS LNA of FIG. 2 does not suffer the gain degradation of the PDC LNA of FIG. 1 due to the first order transconductance signal components of the cancel path canceling some of the first order transconductance signal component output by the main transistor, the CCMDS LNA of FIG. 2 has other drawbacks. One drawback is that, in addition to the gate of a main transistor being coupled to an input lead, there is an additional capacitor that is coupled to the input lead. This extra capacitor that is coupled to the input lead increases the input capacitance of LNA. To interface the LNA to an antenna, an impedance matching network involving an inductor is typically employed. Increasing the input capacitance of the LNA requires that this inductor in the impedance matching network be larger as well. This is undesirable because providing the larger inductor involves increasing the parasitic resistance of the inductor and thus resulting in noise factor degradation.
A second drawback is that the merging nodes 17 and 18 where the cancel and main signals are combined are the output nodes of the LNA of FIG. 2. As the receiver operates, if the impedance of the circuitry that the LNA drives (for example, the mixer in a receive chain) changes, then this impedance change affects cancellation of the third order distortion between the main and cancel paths. This is undesirable.